The present invention relates generally to amplifier driver circuits, particularly to amplifier driver circuits composed of current feedback amplifiers, and more particularly to such amplifier driver circuits which avoid instability in response to both differential input signals and common-mode input signals. The invention also relates to amplifier driver circuits composed of voltage feedback amplifiers, and more particularly to such amplifier driver circuits which avoid instability in response to both differential input signals and common-mode input signals.
Current feedback amplifiers (CFA) are generally considered to be the best candidates for DSL applications for a number of reasons, including their higher slew rates and low inverting-input impedances, to name a few. Most CFAs are made of an input buffer, 2 main current mirrors, and an output stage. In certain amplifier driver applications, such as DSL (Digital Subscriber Line) applications, a pair of current feedback amplifiers are connected together to form a differential amplifier driver circuit. (An equivalent circuit of a current feedback amplifier is shown in FIG. 9)
“Prior Art” FIG. 1 herein illustrates a conventional current feedback amplifier. The gain of differential amplifier driver circuits typically is set to a value of approximately ten. To maximize the bandwidth of differential amplifier driver circuits, the pair of individual amplifiers in such a differential amplifier driver circuit are internally compensated such that they are optimized to operate at these higher gains. Consequently, the individual amplifiers and hence the differential amplifier driver circuits amplifiers composed thereof tend to be unstable if operated at lower gains, and particularly at a gain of unity. This represents a severe problem for the differential amplifier driver circuits when used in DSL applications. In this case, even though the differential amplifier driver circuit is set up for a gain of ten when amplifying differential input signals, it actually operates with a gain of one when amplifying common mode input signals. Because of this, the differential amplifier driver circuit is susceptible to common mode instability.
An alternate compensation method uses AC coupling of each of a pair of individual amplifiers of which a differential amplifier driver circuit is constructed in such a way that the individual amplifiers “see” smaller compensation capacitances during differential input signal amplifying operation and larger compensation capacitances in common mode input signal amplifying operation. U.S. Pat. No. 6,867,649 entitled “Common-Mode and Differential-Mode Compensation for Operational Amplifier Circuits” issued Mar. 15, 2005 to Jeffrey S. Lehto describes the method. “Prior Art” FIG. 2A is a copy of FIG. 4 of U.S. Pat. No. 6,867,649, wherein a compensation scheme is provided for two interconnected amplifier circuits that allows independent frequency compensation of the common-mode and differential signal paths. This method can be used to stabilize differential circuits without compromising performance through over-compensation, and without any need to isolate the amplifiers from one another. However, as a matter of actual circuit implementation, adding an inverting amplifier to each amplifier block is not practical. Specifically, FIG. 2A is a very basic diagram of the circuit, and implementing the illustrated amplifiers 406 and 416 with a gain of −1 is impractical.
“Prior Art” FIG. 2B herein is a copy of FIG. 5 of U.S. Pat. No. 6,867,649, and shows a practical implementation of the invention disclosed therein. This circuit includes two operational amplifier amplifiers AMPA and AMPB. The amplifiers making up AMPA and AMPB can use any operational amplifier topology including voltage feedback and current feedback methods, and can be made from any transistor technology including, but not limited to, bipolar and MOSFET devices. The amplifiers AMPA and AMPB each include a transconductance stage and output buffer, similar to FIG. 1. Circuitry is further included in each of amplifiers AMPA and AMPB to form an inverter, with the inverter having an input connected to the gain node at the output of the transconductance stage.
Common mode compensation is provided by connecting capacitors from the gain node at the input of an inverter in one of the amplifiers AMPA or AMPB to the output of the inverter in the other amplifier. For the bipolar current feedback amplifiers, two capacitors having a value CCOMMON/2 are connected together in each of amplifiers AMPA and AMPB on one end to the output of current mirrors which are connected to effectively form the output of the inverter, and separately to separate inputs of the current mirrors in the opposing amplifiers AMPA or AMPB.
Although the signals of the above mentioned current mirrors are in fact 180 degrees out of phase with one another and thus in a sense form the inputs and outputs of inverters, it is also true that these inputs and outputs are of greatly differing amplitudes. Thus, for differential mode signals, the illustrated “CCOMMON/2” capacitors in fact have substantial changing voltages across them. Since the current through a capacitor is i(t)=C(dv/dt), charging and discharging currents must be supplied to the common mode compensation capacitors during differential mode operation. Because of the symmetry of the connection of the compensation capacitors to the high impedance nodes and to the current mirror inputs, the discharge current of one capacitor is mirrored to become the charge current of another. This effectively makes the common mode compensation capacitors seem almost “invisible” to the high impedance node during differential mode operation, but these currents must flow in the signal current mirrors. This places an undesirable burden on the signal current mirrors which increases with frequency and amplitude. In fact, depending on the magnitude of the charging currents and discharging currents, there is a significant likelihood of high internal AC power dissipation and the possibility of inadequate full power bandwidth of circuits implementing this compensation method.
More specifically, the input signals applied to current mirrors 510 and 562 of FIG. 2B are the inputs to the inverting amplifiers therein. That is, according to the teaching of U.S. Pat. No. 6,867,649, current mirrors 510 and 562 can be thought of as functioning as an inverting amplifiers. However, that actually is incorrect because in current mirror 510, the collector of transistor 521 is held nearly constant at 2 VBE voltage drops below the supply voltage VCC. In the circuit of Prior Art FIG. 2B herein, the collectors of output transistors 503, 504, 553, and 554 have to provide the charging currents and discharging currents to the compensation capacitors CCOMP. The foregoing transistors charge and discharge the compensation capacitors asymmetrically when amplification of the differential input signal occurs in the differential mode. Transistors 503 and 504 (and also transistors 533 and 534) function similarly in either differential mode or common mode operation. One of the current mirrors will be providing additional current beyond what it would ordinarily be provided to charge the large capacitances. However, when the amplification is common mode amplification, the burden of charging and discharging capacitors 514 and 516 (and also capacitors 564 and 566) actually falls entirely on the current mirror output transistors, necessitating the use of substantially larger transistor geometries than would normally be used. For example, if capacitor 514 is being charged, capacitor 516 will be discharging at the same time. These currents will be summed at node VB and be provided from the output transistor of current mirror 562. The current provided by that transistor will be boosted because the opposite capacitors 564 and 566 are discharging into its base so it can provide the charging current to the other set of capacitors 514 and 516. At the same time current mirror 562 is being boosted, current mirror 560 will be turned off because there is an absence of current at the summing node VB. As the frequency increases, larger transistors are needed to supply the current in one current mirror at the same time the other current mirror is being turned off. The above mentioned over-sizing of transistors tends to degrade/distort large signal circuit performance of the amplifier.
The charging current of capacitor 514 will flow into the input of current mirror 510 and also out of its output, while the discharge current of capacitor 516 will cause the current of mirror 512 to decrease. Then the additional current flowing in the output of current mirror 510 will provide the charge and discharge currents of capacitors 564 and 566. These currents in turn cause the current of mirrors 562 and 560 to increase and decrease, respectively. At high amplitudes/frequencies, the discharging capacitors can actually cause the current mirror inputs to be driven below VEE on the negative side and above VCC on the positive side. This, as well as the high charge/discharge current flowing in the complementary current mirror, limits large signal bandwidth, and also results in substantial additional power dissipation. Also, if transistors 503, 504, 553, and 554 are not made sufficiently large, they will have difficulty supplying the charging currents and discharging currents, and therefore those transistors will have a tendency to saturate at high signal levels. This causes the large signal frequency response of the differential amplifier driver circuit in U.S. Pat. No. 6,867,649 to deteriorate rapidly as the signal level increases. Consequently, there is a substantial need for a differential amplifier driver circuit that does not have the foregoing problems.
Differential mode compensation can be provided by connecting a capacitor with value CCOMP/2 from the gain node to ground of each of the amplifiers AMPA or AMPB, similar to the compensation provided in amplifier 100 of FIG. 1. Alternatively, both differential mode and Miller effect compensation can be provided by connecting capacitors from the input to the output of components forming the inverter in each of the amplifiers AMPA and AMPB. For the MOSFET differential amplifiers, differential and Miller effect compensation is provided by connecting a capacitor having a value CCOMP/2 between the inverting and non-inverting outputs in each amplifier. For bipolar current feedback amplifiers, a capacitance of CCOMP/2 is connected between the output of current mirrors forming the gain node, and each current mirror input. As a further alternative, differential and common mode compensation can be provided independently by connecting a capacitor with value CDIFF between the outputs of the inverters of the amplifiers AMPA and AMPB.
A major disadvantage of the foregoing circuit is that the large common mode compensation capacitors must be complementarily charged and discharged for differential mode signals. Although this method functions very well for small signals at moderate frequencies, it suffers from high internal power dissipation as the frequency is increased and it has limited large signal bandwidth capability.
Thus, there is an unmet need for a differential amplifier driver circuit which avoids instability in response to both differential input signals and common-mode input signals and which also has higher full power bandwidth than the closest prior art.
There also is an unmet need for a differential amplifier driver circuit which avoids instability in response to both differential input signals and common-mode input signals and which dissipates less power at high frequencies than the closest prior art.
There also is an unmet need for a differential amplifier driver circuit which avoids the impracticability of the prior art circuitry shown in U.S. Pat. No. 6,867,649 and its problems of high internal power dissipation and inability to provide high full power signal bandwidth.